Two-phase/four-phase modulated comb-shaped spread spectrum communication system

ABSTRACT

When communications are performed between a transmitter  2  and a receiver  3  via a transmission path  4 , the transmitter prepares a basic sequence consisting of two-phase or four-phase tips, and transmits an expanded transmission frame which is constructed by repeatedly arranging one or a plurality of the basic sequence so as to obtain a finite length periodic sequence with a comb-form spectrum and by adding the replica of one or a plurality of tips of a back portion and front portion of the finite length periodic sequence to the outside of the front portion and the outside of the back portion of the finite length periodic sequence; and the receiver demodulates the expanded transmission frame using a matched filter which is matched with the finite length periodic sequence which is the central part of the expanded transmission frame, that is not including the added tips. In other words, the present invention can simplify the construction of communication equipment by using two-phase signals or four-phase signals.

TECHNICAL FIELD

The present invention relates to a two-phase/four-phase modulated spreadspectrum communication system with a comb-form spectrum which isdesigned so as to reduce co-channel interference.

BACKGROUND ART

The present applicant has announced an approximately synchronized CDMAsystem that uses signals consisting of pseudo-periodic sequences as asignal design method which makes it possible to avoid no co-channelinterference.

In this signal design method, interference between desired and undesiredchannels can be eliminated; as a result, the signal of a desired channelcan be efficiently separated from the other undesired channels. However,in conventional approximately synchronized CDMA systems, signals aredesigned by using multi-phase sequences; consequently, the signals onrespective channels are not two-phase signals (which are signalsexpressed by +, −) or four-phase signals (which are signals expressed by+, −, j, −j), and therefore, they should be made of much morecomplicated communication equipment.

The present invention was devised in light of the above facts; and inclaim 1, the object of the present invention is to provide atwo-phase/four-phase modulated spread spectrum communication system witha comb-form spectrum which makes it possible to simplify theconstruction of a communication equipment by means of two-phase signalsor four-phase signals.

Furthermore, in claim 2, the object of the present invention is toprovide a two-phase/four-phase modulated spread spectrum communicationsystem with a comb-form spectrum which makes it possible to assigndifferent carrier waves to respective users by means of two-phasesignals or four-phase signals.

Furthermore, in claim 3, the object of the present invention is toprovide a two-phase/four-phase modulated spread spectrum communicationsystem with a comb-form spectrum which makes it possible, usingtwo-phase signals or four-phase signals, to prevent the generation ofside lobes in the vicinity of the main pulses in signals produced fromthe matched filter on the reception side.

Furthermore, in claim 4, the object of the present invention is toprovide a two-phase/four-phase modulated spread spectrum communicationsystem with a comb-form spectrum which makes it possible to create aplurality of code words by the use of two-phase signals or four-phasesignals.

DISCLOSURE OF INVENTION

In order to accomplish the above-described objects, in claim 1, thetwo-phase/four-phase modulated spread spectrum communication system witha comb-form spectrum of the present invention is characterized by thefact that a basic sequence consisting of two-phase or four-phase chipsis prepared, and an extended transmission frame is constructed byrepeating one or a plurality of the basic sequence so as to obtain aperiodic sequence of finite length with a comb-form spectrum and byduplicating and adding one or a plurality of chips of the back portionand front portion of the periodic sequence of finite length to theoutside of the front portion and outside of the back portion of theperiodic sequence of finite length; and on the reception side theextended transmission frame is demodulated using a matched filter whichis matched with the periodic sequence of finite length prior toextension.

Furthermore, in claim 2, the present invention is characterized by thefact that in the two-phase/four-phase modulated spread spectrumcommunication system with a comb-form modulated spectrum claimed inclaim 1, different carrier waves are assigned to respective users, andtwo-phase signals or four-phase signals with a comb-form modulatedspectrum are assigned to the respective carrier waves thus assigned.

In addition, in claim 3, the present invention is characterized by thefact that in the two-phase/four-phase modulated spread spectrumcommunication system with a comb-form modulated spectrum claimed inclaim 1 or 2, the basic sequence is formed as a two-phase or four-phaseorthogonal sequence or as a multi-phase orthogonal sequence, and sidelobes are prevented from being generated in the vicinity of the mainpulses produced from the matched filter on the reception side, thusreinforcing the anti-multi-path characteristics.

Furthermore, in claim 4, the present invention is characterized by thefact that in the two-phase/four-phase modulated spread spectrumcommunication system with a comb-form modulated spectrum claimed inclaim 1, a plurality of different types of the extended transmissionframe are prepared so as to be used as code words.

BRIEF DESCRIPTION OF DRAWINGS

The above-mentioned features and objects of the present invention willbecome more apparent with reference to the following description takentogether with the accompanying drawings wherein like reference numeralsdenote like elements and in which:

FIG. 1 is a block diagram which illustrates a communication system usingone embodiment of the two-phase/four-phase modulated spread spectrumcommunication system with a comb-form spectrum provided by the presentinvention;

FIGS. 2(a) and 2(b) show examples of the spectrum and waveform of thebasic signal used in the communication system shown in FIG. 1;

FIG. 3 shows an example of the spectrum of the signal obtained byrepeating the signal shown in FIGS. 2(a) and 2(b);

FIGS. 4(a)-4(e) show examples of the spectrum obtained when the signalshown in FIG. 3 is transmitted as a carrier wave at various frequencies;

FIG. 5 shows an example of the spectrum of another basic signal used inthe communication system shown in FIG. 1;

FIG. 6 shows an example of the spectrum of still another basic signalused in the communication system shown in FIG. 1;

FIG. 7 is a model diagram which illustrates one example of a signalwhich acts as the basic signal of a pseudo-periodic signal used in thecommunication method shown in FIG. 1;

FIG. 8 is a model diagram which illustrates an example of apseudo-periodic signal based on the signal shown in FIG. 7; and

FIG. 9 is a model diagram which is used to illustrate the effect of thepseudo-periodic signal shown in FIG. 8.

BEST MODE FOR CARRYING OUT THE INVENTION

First, prior to describing the details of the approximately synchronizedCDMA system of the present invention, the basic technology of thetwo-phase/four-phase modulated spread spectrum communication system witha comb-form spectrum will be described.

<<Preface>>

First, in regards to a signal design method for an approximatelysynchronized CDMA system which can be realized without co-channelinterference, the present inventors have published a publication “N.Suehiro, ‘Approximately synchronized CDMA system without co-channelinterference using pseudo-periodic sequences’, Proceedings ofInternational Symposium on Personal Communications '93—Nanjing, October1993”, a publication “N. Suehiro, ‘A signal design without co-channelinterference for approximately synchronized CDMA systems’, IEEE Journalof Selected Areas in Communications, June 1994”, and a publication “N.Suehiro, ‘Signal design for approximately synchronized CDMA systemswithout co-channel interference’, Proceedings of ISSSTA94, July 1994”.

Furthermore, in regard to a code sequence design method which realizesan excellent data communication capacity in CDMA systems includingapproximately synchronized CDMA systems, the present inventors havepublished a publication “N. Suehiro, ‘Signal design for CDAM by codedaddition of sequences’, Technical Report of IEEE, Vol. IT-, May 1994”, apublication “N. Suehiro, ‘Signal design for approximately synchronizedCDMA systems without co-channel interference’, Proceedings of ISSSTA94,July 1994”, and a publication “N. Suehiro, ‘New signal design method bycoded addition of sequences’, Proceedings of ISIT, September 1995”.

The signals used in these respective publications possess such anextremely important special feature that in cases where a multi-phaseorthogonal sequence is used as the basic orthogonal sequence preparedfor carrying data information, then when a receiver detects it with amatched filter, no side lobe is generated in a small time differencerange of the matched filter output. This special feature will bedescribed in detail below.

First, in terms of the structure of an equipment, a two-phase signal orfour-phase signal is more useful than a multi-phase signal; accordingly,the design method of a two-phase signal or four-phase signal for use inan approximately synchronized CDMA system will be described in whichfeatures of no side lobes in detection and no co-channel interferenceand, excellent data communication capacity will be explained.

In the theory of signal design methods already proposed, a two-phase orfour-phase sequence is used as a means of carrying information in eachset which consists of a transmitter and receiver (generally a basestation). In this case, co-channel interference appears as a multi-phasesequence as far as the carrier means on the receiver is concerned. Inaddition, such a co-channel interference does not affect the sensitivityon the receiver. However, the co-channel interference signal becomes atwo-phase signal or four-phase signal as far as the transmitter carriermeans is concerned.

<<Signal Design Method for Approximately Synchronized CDMA SystemsWithout Co-Channel Interference>>

<Pseudo-Periodic Sequences>

First, prior to a detailed description which will be presented below,the concept of a pseudo-periodic sequence will be outlined.

The correlation characteristics of a period sequence are more easilydesigned than those of a sequence of finite length.

Here,

A=(a₀, a₁, . . . , a_(N−1))

is taken as a sequence of finite length with a length of N which isdesigned so that this sequence has favorable periodic correlationcharacteristics, and

A′=(a_(N−L1), . . . , a_(N−1), a₀, . . . , a_(N−1), a₀, . . . ,a_(L2−1))

is taken as an extended sequence of finite length with a length ofN+L₁+L₂ whose central portion, a portion of length N, coincides with thefinite length sequence A.

In this case, the elements of the front-end portion of the finite lengthsequence A′ with a length of L₁ coincide with the elements of theback-end portion of the finite length sequence A with a length of L₁.Meanwhile, the elements of the back-end portion of the finite lengthsequence A′ with a length of L₂ coincide with the elements of thefront-end portion of the finite length sequence A with a length of L₂.Accordingly, the extended finite length sequence A′ constitutes a finitelength sequence which has a pseudo-period of N.

In a case where the finite length sequence A′ is applied into a filterwhich is matched with the finite length sequence A, the output signalwith length 2N+L₁+L₂−1 that is produced from this filter corresponds tothe correlation function of the finite length sequence A′ and the finitelength sequence A. In addition, the central portion of the output signalwith a length of L₁+L₂+1 also coincides with the portion extending fromthe −L₁ shift component to the L₂ shift component of the autocorrelation function of the periodic sequence “ . . . AAA . . . ”.

Furthermore, in a case where B is a sequence with a finite length of Nwhich differs from that of A, and the finite length sequence A′ isapplied to a filter which is matched with this finite length sequence B,as in the case described above, the central portion of the output signalwith a length of L₁+L₂+1 is produced from the filter coincides with theportion extending from the −L₁ shift component to the L₂ shift componentof the cross correlation function between the two periodic sequences “ .. . AAA . . . ” and “ . . . BBB . . . ”.

Here, the finite length sequence A′ is referred to as a pseudo-periodicsequence with a length of N+L₁+L₂.

<Design of Periodic Sequences Which Do Not Have Correlation>

The present inventors have proposed a method for setting, periodicsequences which do not have correlation. Here, this design method willagain be explained. First, let A and B be cyclic matrices, each of whichrepresents a periodic sequence, then these matrices A and B are relatedto diagonal matrices C and D as indicated by the following equations:

A=F⁻¹CF

 B=F⁻¹DF

Here, F indicates a DFT (Discrete Fourier transform) matrix.

The correlation function between the periodic sequences expressed by thecyclic matrices A and B can be expressed by the following equation:

Numerical formula 1

A ^(t){overscore (B)}=F⁻¹C{overscore (D)}F  (1)

Here, {overscore (B)} and {overscore (D)} are respective complexconjugates of B and D, and ^(t){overscore (B)} is a transposed matrix of{overscore (B)}. Furthermore, in a case where all of the diagonalelements in the diagonal matrices on the right side of theabove-described Equation (1), i. e.,

Numerical formula 2

Diagonal matrices C, {overscore (D)}

are zero, the cross correlation function between the cyclic matrices Aand B is zero in all terms.

For example, since (1, 1, W₃) is an orthogonal sequence, each of thecolumn vectors on the right side of Equation (2) is an orthogonalsequence. Transforming an orthogonal sequence with DFT makes a(multi-phase) sequence whose components have a certain absolute value.Accordingly, $\begin{matrix}{{Numerical}\quad {formula}\quad 3} & \quad \\{{F_{12}^{- 1}\begin{bmatrix}1 & 0 & 0 & 0 \\0 & 1 & 0 & 0 \\0 & 0 & 1 & 0 \\0 & 0 & 0 & 1 \\1 & 0 & 0 & 0 \\0 & 1 & 0 & 0 \\0 & 0 & 1 & 0 \\0 & 0 & 0 & 1 \\W_{3} & 0 & 0 & 0 \\0 & W_{3} & 0 & 0 \\0 & 0 & W_{3} & 0 \\0 & 0 & 0 & W_{3}\end{bmatrix}} = {\frac{1}{2}\begin{bmatrix}W_{12}^{1} & W_{12}^{1} & W_{12}^{1} & W_{12}^{1} \\W_{12}^{1} & W_{12}^{2} & W_{12}^{3} & W_{12}^{4} \\W_{12}^{9} & W_{12}^{11} & W_{12}^{1} & W_{12}^{3} \\W_{12}^{1} & W_{12}^{4} & W_{12}^{7} & W_{12}^{10} \\W_{12}^{1} & W_{12}^{5} & W_{12}^{9} & W_{12}^{1} \\W_{12}^{9} & W_{12}^{2} & W_{12}^{7} & W_{12}^{0} \\W_{12}^{1} & W_{12}^{7} & W_{12}^{1} & W_{12}^{7} \\W_{12}^{1} & W_{12}^{8} & W_{12}^{3} & W_{12}^{10} \\W_{12}^{9} & W_{12}^{5} & W_{12}^{1} & W_{12}^{9} \\W_{12}^{1} & W_{12}^{10} & W_{12}^{7} & W_{12}^{4} \\W_{12}^{1} & W_{12}^{11} & W_{12}^{9} & W_{12}^{7} \\W_{12}^{9} & W_{12}^{6} & W_{12}^{7} & W_{12}^{6}\end{bmatrix}}} & (2)\end{matrix}$

Here,$W_{N} = {\exp \left( \frac{2\quad \pi \sqrt{- 1}}{N} \right)}$

F: 12-point DFT matrix.

Four multi-phase periodic sequences can be obtained by using the columnsof the right-side matrix of the above Equation (2). All of the terms ofthe cross-correlation functions between any pair of these fourmulti-phase periodic sequences become zero; accordingly, the results ofmultiplying the corresponding terms in these spectra take zero in allterms.

<<Design of Set of Periodic Sequences Which Do Not HaveCross-Correlation or Auto Correlation Side Lobes in the Case of VerySmall Shifts>>

Furthermore, since (1, 1, W₃) is an orthogonal sequence, each of thefour columns in the left-side matrix of Equation (2) is an orthogonalsequence.

The auto correlation functions of these four-phase periodic sequencesare as follows:

Numerical formula 4

(1, 0, 0, 1, 0, 0, 1, 0, 0, 1, 0, 0)

(1, 0, 0, j, 0, 0, −1, 0, 0, −j, 0, 0)

(1, 0, 0, −1, 0, 0, 1, 0, 0, −1, 0, 0)

(1, 0, 0, −j, 0, 0, −1, 0, 0, j, 0, 0)

Here, j={square root over (−1+L )}

Furthermore, when they are used as pseudo-periodic signals, these autocorrelation functions have useful characteristics for separatingreceived signal and multi-path noise.

This point will be described in detail below. First, let (a₀, a₁, a₂) bea multi-phase sequence with a period of 3, and (c₀, c₁, c₂) be anorthogonal sequence with a period of 3, then $\begin{matrix}{{Numerical}\quad {formula}\quad 5} \\{{F_{3}\begin{bmatrix}a_{0} \\a_{1} \\a_{2}\end{bmatrix}} = \begin{bmatrix}c_{0} \\c_{1} \\c_{2}\end{bmatrix}}\end{matrix}$

is obtained. And the following two equations, $\begin{matrix}\begin{matrix}{{Numerical}\quad {formula}\quad 6} \\{{\sqrt{2}{F_{6}^{- 1}\begin{bmatrix}c_{0} \\0 \\c_{1} \\0 \\c_{2} \\0\end{bmatrix}}} = {\begin{bmatrix}a_{0} \\a_{1} \\a_{2} \\a_{0} \\a_{1} \\a_{2}\end{bmatrix}\quad {and}}}\end{matrix} \\{{Numerical}\quad {formula}\quad 7} \\{{\sqrt{2}{F_{6}^{- 1}\begin{bmatrix}0 \\c_{0} \\0 \\c_{1} \\\begin{matrix}0 \\c_{2}\end{matrix}\end{bmatrix}}} = \begin{bmatrix}{w_{6}^{0}a_{0}} \\{w_{6}^{1}a_{1}} \\{w_{6}^{2}a_{2}} \\{w_{6}^{3}a_{0}} \\{w_{6}^{4}a_{1}} \\{w_{6}^{5}a_{2}}\end{bmatrix}}\end{matrix}$

are compared, then, in a case where the auto correlation function of(a₀, a₁, a₂, a₀, a₁, a₂) is expressed as $\begin{matrix}{{Numerical}\quad {formula}\quad 8} \\{{F_{6}^{- 1}\begin{bmatrix}{c_{0}\overset{\_}{c_{0}}} \\0 \\{c_{1}\overset{\_}{c_{1}}} \\\begin{matrix}0 \\{c_{2}\overset{\_}{c_{2}}} \\0\end{matrix}\end{bmatrix}} = \begin{bmatrix}e_{0} \\e_{1} \\e_{2} \\e_{0} \\e_{1} \\e_{2}\end{bmatrix}}\end{matrix}$

it can be seen that the auto correlation function of

(w₆ ⁰a₀, w₆ ¹a₁, w₆ ²a₂, w₆ ³a₀, w₆ ⁵a₁,w₆ ⁵a₂)

is as follows $\begin{matrix}{{Numerical}\quad {formula}\quad 9} \\{{F_{6}^{- 1}\begin{bmatrix}0 \\{c_{0}\overset{\_}{c_{0}}} \\0 \\{c_{1}\overset{\_}{c_{1}}} \\\begin{matrix}0 \\{c_{2}\overset{\_}{c_{2}}}\end{matrix}\end{bmatrix}} = \begin{bmatrix}{w_{6}^{0}e_{0}} \\{w_{6}^{1}e_{1}} \\{w_{6}^{2}e_{2}} \\{w_{6}^{3}e_{0}} \\{w_{6}^{4}e_{1}} \\{w_{6}^{5}e_{2}}\end{bmatrix}}\end{matrix}$

Here, if e₁=e₂=0, then $\begin{matrix}{{Numerical}\quad {formula}\quad 10} \\{\begin{bmatrix}{c_{0}\overset{\_}{c_{0}}} \\0 \\{c_{1}\overset{\_}{c_{1}}} \\\begin{matrix}0 \\{c_{2}\overset{\_}{c_{2}}} \\0\end{matrix}\end{bmatrix} = {{F_{6}\begin{bmatrix}e_{0} \\0 \\0 \\e_{0} \\0 \\0\end{bmatrix}} = {{\frac{1}{\sqrt{6}}\begin{bmatrix}{2e_{0}} \\0 \\{2e_{0}} \\0 \\{2e_{0}} \\0\end{bmatrix}}\quad {and}}}} \\{{Numerical}\quad {formula}\quad 11} \\{\begin{bmatrix}0 \\{c_{0}\overset{\_}{c_{0}}} \\0 \\{c_{1}\overset{\_}{c_{1}}} \\\begin{matrix}0 \\{c_{2}\overset{\_}{c_{2}}}\end{matrix}\end{bmatrix} = {{F_{6}\begin{bmatrix}e_{0} \\0 \\0 \\{- e_{0}} \\0 \\0\end{bmatrix}} = {\frac{1}{\sqrt{6}}\begin{bmatrix}0 \\{2e_{0}} \\0 \\{2e_{0}} \\0 \\{2e_{0}}\end{bmatrix}}}}\end{matrix}$

Consequently, it is evident that when e₁=e₂=0,

|c₀|=|c₁|=|c₂|

This means that when

(c₀, c₁, c₂)

is a multi-phase orthogonal sequence, the auto correlation function of

(a₀, a₁, a₂, a₀, a₁, a₂)

and the auto correlation function of

(w₆ ⁰a₀, w₆ ¹a₁, w₆ ²a₂, w₆ ³a₀, w₆ ⁵a₁,w₆ ⁵a₂)

have no side lobes at the shifts of −2, −1, 1 or 2.

This conclusion can easily be generalized.

<<Approximately Synchronized CDMA System Which Has No Side Lobes inDetection or Co-Channel Interference>>

Furthermore, since (1, 1, W₃) is a multi-phase orthogonal sequence inEquation (2), the pseudo-periodic sequence A_(i)′ has no side lobes atthe shifts of −2, −1, 1 or 2 in a case where (1, 1, W₃) is applied to amatched filter matched to A_(i) (i=1, 2, 3 or 4). Here, the columnvectors on the right side of Equation (2) are taken as A1, A2, A3 andA4.

Furthermore, in a case where another pseudo-periodic sequence A_(j)′with a length of 12+2L and a pseudo-period of 12 is applied to a matchedfilter matched to A_(i), the output signal produced from the filterbecomes zero from the −L shift period to the L shift period.Accordingly, an approximately synchronized state which is such that thedifference between the signal and the interference is within the rangeextending from −L to L is produced, so that a CDMA communication systemwhich has no side lobes in detection and no co-channel interference canbe realized in the shift ranging from −L to L.

<<Two-Phase or Four-Phase Signal Design Method for ApproximatelySynchronized CDMA System with No Side Lobes in Detection or Co-ChannelInterference>>

By using a DFT matrix, it is possible to convert an orthogonal sequenceinto a multi-phase periodic sequence; and by using the above-describedDFT matrix, it is possible to convert a multi-phase periodic sequenceinto an orthogonal sequence.

Accordingly, by using the DFT matrix, it is possible to convert onemulti-phase orthogonal sequence into another multi-phase orthogonalsequence.

Here, since (1, j) is an orthogonal sequence, H1 shown in the followingequation is a cyclic type unitary matrix. $\begin{matrix}{{Numerical}\quad {formula}\quad 12} \\{H_{1} = {\frac{1}{\sqrt{2}}\begin{bmatrix}1 & j \\j & 1\end{bmatrix}}}\end{matrix}$

Likewise, the eight columns in the respective matrices shown belowconstitute a super-regular polyhedron in Euclidean space.$\begin{matrix}{{Numerical}\quad {formula}\quad 13} \\{{\sqrt{2}H_{1}} = \begin{bmatrix}1 & j \\j & 1\end{bmatrix}} \\{{j\sqrt{2}H_{1}} = \begin{bmatrix}j & {- 1} \\{- 1} & j\end{bmatrix}} \\{{{- \sqrt{2}}H_{1}} = \begin{bmatrix}{- 1} & {- j} \\{- j} & {- 1}\end{bmatrix}} \\{{{- j}\sqrt{2}H_{1}} = \begin{bmatrix}{- j} & 1 \\1 & {- j}\end{bmatrix}}\end{matrix}$

Here, since the respective columns in $\begin{matrix}{{Numerical}\quad {formula}\quad 14} \\{{\sqrt{2}F_{2}H_{1}} = \begin{bmatrix}W_{8}^{1} & W_{8}^{1} \\W_{8}^{7} & W_{8}^{3}\end{bmatrix}} \\{{j\sqrt{2}F_{2}H_{1}} = \begin{bmatrix}W_{8}^{3} & W_{8}^{3} \\W_{8}^{1} & W_{8}^{5}\end{bmatrix}} \\{{{- \sqrt{2}}F_{2}H_{1}} = \begin{bmatrix}W_{8}^{5} & W_{8}^{5} \\W_{8}^{3} & W_{8}^{7}\end{bmatrix}} \\{{{- j}\sqrt{2}F_{2}H_{1}} = \begin{bmatrix}W_{8}^{7} & W_{8}^{7} \\W_{8}^{5} & W_{8}^{1}\end{bmatrix}}\end{matrix}$

are orthogonal sequences, eight four-phase code words including asuper-regular polyhedron in signal space can be prepared for respectiveusers as shown in the following equations: $\begin{matrix}{{Numerical}\quad {formula}\quad 15} \\\begin{matrix}{{\sqrt{3}{F_{6}^{- 1}\begin{bmatrix}w_{8}^{1} & w_{8}^{1} & w_{8}^{3} & w_{8}^{3} & w_{8}^{5} & w_{8}^{5} & w_{8}^{7} & w_{8}^{7} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\w_{8}^{7} & w_{8}^{3} & w_{8}^{1} & w_{8}^{5} & w_{8}^{3} & w_{8}^{7} & w_{8}^{5} & w_{8}^{1} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0\end{bmatrix}}} = \quad \begin{bmatrix}1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & {- 1} \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j}\end{bmatrix}} \\{= \quad \left\lbrack \begin{matrix}\begin{matrix}y_{00} & y_{01} & y_{02} & y_{03} & y_{04} & y_{05}\end{matrix} & \left. \begin{matrix}y_{06} & y_{07}\end{matrix} \right\rbrack\end{matrix} \right.}\end{matrix} \\{{Numerical}\quad {formula}\quad 16} \\\begin{matrix}{{\sqrt{3}{F_{6}^{- 1}\begin{bmatrix}0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\w_{8}^{1} & w_{8}^{1} & w_{8}^{3} & w_{8}^{3} & w_{8}^{5} & w_{8}^{5} & w_{8}^{7} & w_{8}^{7} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\w_{8}^{7} & w_{8}^{3} & w_{8}^{1} & w_{8}^{5} & w_{8}^{3} & w_{8}^{7} & w_{8}^{5} & w_{8}^{1} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0\end{bmatrix}}} = \quad \begin{bmatrix}w_{24}^{0} & 0 & 0 & 0 & 0 & 0 \\0 & w_{24}^{4} & 0 & 0 & 0 & 0 \\0 & 0 & w_{24}^{8} & 0 & 0 & 0 \\0 & 0 & 0 & w_{24}^{12} & 0 & 0 \\0 & 0 & 0 & 0 & w_{24}^{16} & 0 \\0 & 0 & 0 & 0 & 0 & w_{24}^{20}\end{bmatrix}} \\{\quad \begin{bmatrix}1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & {- 1} \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j}\end{bmatrix}} \\{= \quad \left\lbrack \begin{matrix}\begin{matrix}y_{10} & y_{11} & y_{12} & y_{13} & y_{14} & y_{15}\end{matrix} & \left. \begin{matrix}y_{16} & y_{17}\end{matrix} \right\rbrack\end{matrix} \right.}\end{matrix} \\{{Numerical}\quad {formula}\quad 17} \\\begin{matrix}{{\sqrt{3}{F_{6}^{- 1}\begin{bmatrix}0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\w_{8}^{1} & w_{8}^{1} & w_{8}^{3} & w_{8}^{3} & w_{8}^{5} & w_{8}^{5} & w_{8}^{7} & w_{8}^{7} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\w_{8}^{7} & w_{8}^{3} & w_{8}^{1} & w_{8}^{5} & w_{8}^{3} & w_{8}^{7} & w_{8}^{5} & w_{8}^{1}\end{bmatrix}}} = \quad \begin{bmatrix}w_{24}^{0} & 0 & 0 & 0 & 0 & 0 \\0 & w_{24}^{8} & 0 & 0 & 0 & 0 \\0 & 0 & w_{24}^{16} & 0 & 0 & 0 \\0 & 0 & 0 & w_{24}^{0} & 0 & 0 \\0 & 0 & 0 & 0 & w_{24}^{8} & 0 \\0 & 0 & 0 & 0 & 0 & w_{24}^{16}\end{bmatrix}} \\{\quad \begin{bmatrix}1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j} \\1 & j & j & {- 1} & {- 1} & {- j} & {- j} & 1 \\j & 1 & {- 1} & j & {- j} & {- 1} & 1 & {- j}\end{bmatrix}} \\{= \quad \left\lbrack \begin{matrix}\begin{matrix}y_{20} & y_{21} & y_{22} & y_{23} & y_{24} & y_{25}\end{matrix} & \left. \begin{matrix}y_{26} & y_{27}\end{matrix} \right\rbrack\end{matrix} \right.}\end{matrix}\end{matrix}$

In this way, it can be seen that each pseudo-periodic signal set[y_(i0′)y_(i1′), . . . , y_(i7′)] is a four-phase signal set its owncarrier wave.

Meanwhile, in cases where the entire system is approximatelysynchronized, signals from undesired transmitters are received asmulti-phase signals which do not produce any filter output at thereceiver; accordingly, such signals do not produce co-channelinterference.

Furthermore, in order to design four-phase signals, four-phaseorthogonal sequences such as (1, 1, 1, −1), (1, j, 1, −j), (1, 1, 1, 1,1, j, −1, −j, 1, −1, 1, −1, 1, −j, −1, j) or (1, 1, 1, 1, 1, j, −1, j,1, −1, 1, −1, 1, j, −1, j), etc. can be used instead of (1, j).

Moreover, a two-phase orthogonal sequence (1, 1, 1, −1) can be used inorder to set two-phase signals.

<<Design of Two-Phase or Four-Phase Signals for ApproximatelySynchronized CDMA System Which Does Not Have Co-Channel Interference,But Which Does Have Side Lobes in Detection>>

Furthermore, in cases where side lobes in detection are permitted,arbitrary two-phase or four-phase codes can be used.

If $\begin{matrix}{{Numerical}\quad {formula}\quad 18} \\{X_{i} = \begin{bmatrix}X_{i,o} \\X_{i,o} \\\vdots \\X_{i,{N - 1}}\end{bmatrix}}\end{matrix}$

is taken as a code word in a two-phase or four-phase code of length N,then the orthogonal sequence F_(N) x_(i) can be used as the basicorthogonal sequence in the above-described Equation (2).

In this way, each of the signals obtained is a two-phase or four-phasesignal on its own carrier wave. In other words, any arbitrarily selectedconventional codes can be used for the signal design described so far.

<<π/4 Rotation for Four-Phase Signals>>

Furthermore, by rotating four-phase signals by π/4 on the complex plane,balanced four-phase signals containing $\begin{matrix}{{Numerical}\quad {formula}\quad 19} \\\left( {\frac{1 + j}{\sqrt{2}},\frac{{- 1} + j}{\sqrt{2}},\frac{{- 1} - j}{\sqrt{2}},\frac{1 - j}{\sqrt{2}}} \right)\end{matrix}$

can be obtained from unbalanced four-phase signals containing (1, j, −1,−j).

In some cases, the balanced four-phase signals may be more suitable thanthe unbalanced four-phase signals from the standpoint of theconstruction of the equipment. For example, since the real-number partsand imaginary-number parts of the balanced four-phase signalsrespectively consist of two values each, it may be possible to simplifythe construction of the equipment depending on the approach used.

Hereafter, the two-phase/four-phase modulated spread spectrumcommunication system with a comb-form spectrum provided by the presentinvention, which uses the basic technique described above, will bedescribed in detail with reference to the attached drawings.

FIG. 1 is a block diagram which illustrates one example of acommunication system using one embodiment of the two-phase/four-phasemodulated spread spectrum communication system with a comb-form spectrumprovided by the present invention.

In the communication system 1 shown in this figure, when communicationis to be performed between a transmitter 2 and a receiver 3 via atransmission path 4, a basic sequence consisting of two-phase orfour-phase chips is prepared on the transmission side, and an extendedtransmission frame is constructed by repeatedly arranging the basicsequence so as to make a finite length periodic sequence, and adding oneor a plurality of chips of the back portion and front portion of thefinite length periodic sequence with a comb-form spectrum, to theoutside of the front portion and the outside of the back portion of thefinite length periodic sequence; and at the receiver, this extendedtransmission frame is demodulated with a matched filter matched to thefinite length periodic sequence prior to extension. Furthermore, in thiscase, when two-phase signal or four-phase signals are transmitted andreceived using a carrier wave of a predetermined frequency, i.e., acarrier wave that has a frequency which the transmitter and receiverhave themselves decided to use, even if users who transmit and receiveother carrier waves should receive the two-phase signals or four-phasesignals on such a carrier wave, these signals can be viewed by the usersthemselves as “multi-phase interference signals producing no co-channelinterference under conditions of approximate synchronization” despitethe fact that the signals on other carrier waves are two-phase signalsor four-phase signals; accordingly, two-phase signals or four-phasesignals on respective carrier waves can be assigned to respective users.

Hereafter, the carrier wave and the two-phase signals (or four-phasesignals) used in the above communication system 1 will be described indetail.

Now, assuming that a signal S₁ with modulated spread spectrum exists asa signal designed by the method described above and constructed onlyfrom main pulses which do not have side lobes as shown in FIGS. 2(a) and2(b), if a signal “S₁S₁S₁S₁S₁” is produced by repeating this signal S₁five times, the spectrum of this signal “S₁S₁,S₁S₁S₁” will have acomb-form shape that has a blank-portion width of “4” with respect toeach spectral width of “1”, as shown in FIG. 3.

In this case, as is clear from FIG. 3, if the length of the signal S₁ is8, then the number of pulses will be “8”; accordingly, if this signal“S₁S₁S₁S₁S₁” is carried on five carrier waves of slightly differentfrequencies, the spectra of the respective signals will not overlap witheach other as shown in FIGS. 4(a) through 4(e).

Furthermore, let the signal S₁ be a four-phase signal, it is easy toproduce a four-phase transmission signal on the sender's side;accordingly, the construction of the transmitter 2 can be simplified,and the manufacturing cost can be reduced compared to the system usingpoly-phase signals.

In particular, if a four-phase signal is formed in which the respectivesignal phases are shifted 45 degrees each as shown by the followingequation, both the real part and the imaginary part will consist of thetwo values $\begin{matrix}{{Numerical}\quad {formula}\quad 20} & \quad \\\left( {\frac{1}{\sqrt{2}},\frac{- 1}{\sqrt{2}}} \right) & (3)\end{matrix}$

so that a signal can easily be generated with the following components,$\begin{matrix}{{Numerical}\quad {formula}\quad 21} & \quad \\\left( {\frac{1 + j}{\sqrt{2}},\frac{1 - j}{\sqrt{2}},\frac{{- 1} + j}{\sqrt{2}},\frac{{- 1} - j}{\sqrt{2}}} \right) & (4)\end{matrix}$

Let consider another signal as expressed by the following equation.$\begin{matrix}{{Numerical}\quad {formula}\quad 22} & \quad \\{Y_{00} = \begin{bmatrix}1 \\j \\1 \\j \\1 \\j\end{bmatrix}} & (5)\end{matrix}$

Since Y₀₀ is constructed by repeating a signal (1, j) with a flatspectrum three times; the resulting spectrum is as shown in FIG. 5.Furthermore, in the case of the signal expressed by the followingequation: $\begin{matrix}{{Numerical}\quad {formula}\quad 23} & \quad \\{Y_{10} = {{\begin{bmatrix}W_{24}^{0} & \quad & \quad & \cdots & 0 & 0 \\\quad & W_{24}^{4} & \quad & \quad & \cdots & 0 \\\quad & \quad & W_{24}^{8} & \quad & \quad & \quad \\\quad & \quad & \quad & W_{24}^{12} & \quad & \quad \\0 & \cdots & \quad & \quad & W_{24}^{16} & \quad \\0 & 0 & \cdots & \quad & \quad & W_{24}^{20}\end{bmatrix}\begin{bmatrix}1 \\j \\1 \\j \\1 \\j\end{bmatrix}} = \begin{bmatrix}W_{24}^{0} \\{W_{24}^{4}j} \\W_{24}^{8} \\{W_{24}^{12}j} \\W_{24}^{16} \\{W_{24}^{20}j}\end{bmatrix}}} & (6)\end{matrix}$

Here, W₂₄:W₂₄=expt${W_{24}\text{:}W_{24}} = {\exp \left( \frac{2\quad \pi \sqrt{- 1}}{24} \right)}$

the spectrum is as shown in FIG. 6.

Furthermore, if [Y₀₀, Y₀₁, Y₀₂, Y₀₃, Y₀₄, Y₀₅, Y₀₆ , Y₀₇] is assigned toone user where each component signal is given as follows,$\begin{matrix}{{Numerical}\quad {formula}\quad 24} & \quad \\{Y_{01} = \begin{bmatrix}j \\{- 1} \\j \\{- 1} \\j \\{- 1}\end{bmatrix}} & (7) \\{Y_{02} = \begin{bmatrix}{- 1} \\{- j} \\{- 1} \\{- j} \\{- 1} \\{- j}\end{bmatrix}} & (8) \\{{Numerical}\quad {formula}\quad 25} & \quad \\{Y_{03} = \begin{bmatrix}{- j} \\1 \\{- j} \\1 \\{- j} \\1\end{bmatrix}} & (9) \\{Y_{04} = \begin{bmatrix}j \\1 \\j \\1 \\j \\1\end{bmatrix}} & (10) \\{{Numerical}\quad {formula}\quad 26} & \quad \\{Y_{05} = \begin{bmatrix}{- 1} \\j \\{- 1} \\j \\{- 1} \\j\end{bmatrix}} & (11) \\{Y_{06} = \begin{bmatrix}{- j} \\{- 1} \\{- j} \\{- 1} \\{- j} \\{- 1}\end{bmatrix}} & (12) \\{{Numerical}\quad {formula}\quad 27} & \quad \\{Y_{07} = \begin{bmatrix}1 \\{- j} \\1 \\{- j} \\1 \\{- j}\end{bmatrix}} & (13)\end{matrix}$

then 3 bits of communications can be performed at one time, merely bydetecting whether any of Y₀₀ through Y₀₇ has been transmitted. In thiscase, Y₀₀ through Y₀₇ form an ideal code word constituting asuper-regular polyhedron in the communication space. Furthermore, if[Y₁₀, Y₁₁, Y₁₂, Y₁₃, Y₁₄, Y₁₅, Y₁₆, Y₁₇] as produced by the followingequation is assigned to another user, $\begin{matrix}{{Numerical}\quad {formula}\quad 28} & \quad \\\begin{matrix}{\begin{matrix}\left\lbrack Y_{10} \right. & Y_{11} & Y_{12} & Y_{13} & Y_{14} & Y_{15} & \left. \begin{matrix}Y_{16} & Y_{17}\end{matrix} \right\rbrack\end{matrix} = \quad \begin{bmatrix}W_{24}^{0} & \quad & \quad & \cdots & 0 & 0 \\\quad & W_{24}^{4} & \quad & \quad & \cdots & 0 \\\quad & \quad & W_{24}^{8} & \quad & \quad & \quad \\\quad & \quad & \quad & W_{24}^{12} & \quad & \quad \\0 & \cdots & \quad & \quad & W_{24}^{16} & \quad \\0 & 0 & \cdots & \quad & \quad & W_{24}^{20}\end{bmatrix}} \\{\quad \left\lbrack \begin{matrix}\begin{matrix}Y_{00} & Y_{01} & Y_{02} & Y_{03} & Y_{04} & Y_{05}\end{matrix} & \left. \begin{matrix}Y_{06} & Y_{07}\end{matrix} \right\rbrack\end{matrix} \right.}\end{matrix} & (14)\end{matrix}$

then this user can be allowed to perform communications three bits at atime in the same manner as the previous user, without causing anyspectral interference with the previous user. In this case both usersare using four-phase signals on respective carrier waves.

Thus, with the use of the communication system described above and theconcept of a “pseudo-periodic sequence” that is a concept wherein abasic sequence (for instance signal S₁) consisting of two-phase orfour-phase chips is prepared, and an extended transmission frame isconstructed by repeating one or a plurality of the basic sequence so asto obtain a finite length periodic sequence with a comb-form spectrumand by adding one or a plurality of chips of the back portion and frontportion of the finite length periodic sequence to the outside of thefront portion and outside of the back portion of the finite lengthperiodic sequence, and at the receiver the extended transmission frameis demodulated with a matched filter matched to the finite lengthperiodic sequence prior to extension; it is possible to realize anapproximately synchronized interference-free CDMA system by creating asignal A′ shown in FIG. 8 based upon the signal A of FIG. 7, and bytransmitting this signal A′, and then by demodulating this signal at thereceiver with a matched filter matched to the signal A.

In this case, for example, if a signal A′ as shown in FIG. 9 is producedand transmitted by the transmitter 2, this signal A′ will act as aninterference signal with respect to a receiver 3 who attempts to receivesignal B, e.g., a general receiving station [B] other than receivingstation [A]. The phase at which the receiving station [B] receivessignal A constantly fluctuates. However, as shown in FIG. 9, assumingthat the phase difference between the phase of the demodulating frame ofreceiving station [B] and the phase of the received signal A is equal toor less than −τ₁ to +τ₂ seconds, and that one portion A of signal A′ ismatched to the phase of the despreading sequence of receiving station[B]; then, when the receiving station [B] demodulates signal A″, signalA″ is not generated as an interference component in the matched filteroutput of the receiving station [B], because signal A″, when viewed fromthe receiving station B, is a signal obtained by extracting one periodof the periodic sequence of signal A and the despreading sequence B ofthe receiving station [B] and signal A are orthogonal each other.

Meanwhile, the receiving station [A] detects only signal A by using amatched filter matched to signal A; and even though when the station [A]receives the portion B″ of signal B with the same frame format addressedto the receiving station [B], this does not appear as the output of thematched filter matched to the periodic sequence A. In this way, anapproximately synchronized interference-free CDMA system can berealized.

Thus, in this embodiment, when communications are performed between thetransmitter 2 and the receiver 3 via the transmission path 4, a basicsequence consisting of two-phase or four-phase chips is prepared at thetransmitter, and an extended transmission frame is constructed byrepeatedly arranging one or a plurality of the basic sequence to make afinite length periodic sequence with a comb-form spectrum and by addingone or a plurality of chips of the back portion and front portion of thefinite length periodic sequence to the outside of the front portion andoutside of the back portion of the finite length periodic sequence, andat the receiver side the extended transmission frame is demodulated witha matched filter matched to the finite length periodic sequence prior toextension; accordingly, the construction of the communication equipmentcan be simplified, because a two-phase signal or four-phase signal isused, compared to cases where multi-phase signals are used, and also themanufacturing costs can be reduced.

Furthermore, in this embodiment, a transmitter produces the signal A′ onthe basis of signal A, which is the object of transmission, andtransmits this signal, and receives, demodulates the received signalwith a matched filter matched to the signal A, using the above-describedcommunication method and the concept of a “pseudo-periodic sequence”, anapproximately synchronized interference-free CDMA system can berealized.

In addition, in the embodiment described above, communications areperformed using the four-phase signals shown in Equation (3), etc.;however, it is also possible to perform communications using thetwo-phase signal shown in the following formula:

(1, j)  (15)

or the four-phase signals shown in the formula below, etc.$\begin{matrix}{{Numerical}\quad {formula}\quad 29} & \quad \\\left. \begin{matrix}\left( {1,1,1,{- 1}} \right) \\\left( {1,1,1,1,1,j,{- 1},{- j},1,{- 1},1,{- 1},1,{- j},{- 1},j} \right) \\\left( {1,1,1,1,1,{- j},{- 1},j,1,{- 1},1,{- 1},1,j,{- 1},{- j}} \right)\end{matrix} \right\} & (16)\end{matrix}$

Furthermore, the present inventors have published a method for producingmulti-phase signals which satisfy the above-described Numerical Formula29 in Modulatable Orthogonal Sequences and their Application to SSAMSystems (IEEE Transactions on Information Theory, Vol. IT-34, No.1,January 1988). The four-phase signals produced using this method ofproduction are as follows:

(1) 0000012302020321

(1′) 0000032102020123

(2) 0002012102000323

(2′) 0002032302000121

(3) 0010013302120331

(3′) 0030031102320113

(4) 0001123120213211

(4′) 0003321320231233

(5) 0002123220223212

(5′) 0002321220221232

(6) 0022010102200303

(6′) 0022030302200101

(7) 0011302320311003

(7′) 0033102120133001

(8) 0003123320233213

(8′) 0001321120211231

(9) 0001012002030322

(9′) 0003032002010122

(9″) 0002230302002101

(9′″) 0002210102002303

(10) 0000301220201032

(10′) 0000103220203012

(10″) 0000230102022103

(10′″) 0000210302022301

(11) 0011013002130332

(11′) 0033031002310112

(11″) 001023112122113

(11′″) 0030213302322331

(12) 0001301320211033

(12′) 0003103120233011

(12″) 003301120231031

(12′″) 0001103320213013

Thus, a total of 32 different types of four-phase signals can beobtained.

Here, (0, 1, 2, 3) corresponds to (exp(ja), exp(j(a+π/2)), exp(j(a+π)),exp(j(a+3π/z). In this case, pi is π is the ratio of the circumferenceof a circle to its diameter; and the above situation occurs when a=π/4.

In this case as well, the spectra of these two-phase signals orfour-phase signals are flat; and therefore, an effect similar to that ofthe embodiment described above can be obtained.

Furthermore, in the above embodiment, a signal “S₁S₁S₁S₁S₁” is producedby repeating a signal S₁ of an orthogonal sequence in order to realizean interference-free CDMA system. However, if the signal S_(i) isrepeated, the resulting spectrum is a comb-form spectrum, and as aresult, it is also possible to produce a transmission signal byrepeating a basic sequence signal other than such an orthogonalsequence.

Furthermore, in the embodiment above, each component of the comb-formspectrum takes the same amplitude by using the orthogonal sequencesignal S₁, thus insuring desirable properties as a signal fortransmission and reception. However, as a signal S₁, it is also possibleto use a multi-phase orthogonal sequence (which is an orthogonalsequence with a certain absolute value) in stead of such an ordinaryorthogonal sequence, so that the side lobes in the vicinity of the mainpulses may become zero when the received signal is correlativelydetected at the receiver 3.

By using such an arrangement, it is possible to facilitate thediscrimination between multi-path pulses and main pulses, thus improvingthe anti-multi-path characteristics, even in cases where signalstransmitted from the transmitter 2 are received by a receiver via aplurality of paths.

As described above, according to the present invention, in claim 1, theconstruction of the communication equipment can be simplified as aresult of the use of two-phase signals or four-phase signals.

Furthermore, in claim 2, different carrier waves can be assigned torespective users by using two-phase signals or four-phase signals.

Furthermore, in claim 3, the occurrence of side lobes in the vicinity ofthe main pulses in the matched filter output at the receiver can beprevented by using two-phase signals or four-phase signals.

Furthermore, in claim 4, a plurality of code words can be created byusing two-phase signals or four-phase signals.

What is claimed is:
 1. A process for two-phase/four-phase modulatedspread spectrum communication using a comb-form spectrum, said processcomprising: preparing a basic sequence consisting of two-phase orfour-phase chips, transmitting with a transmitter an extendedtransmission frame which is constructed by repeatedly arranging one or aplurality of said basic sequence so as to obtain a finite lengthperiodic sequence with a comb-form spectrum, adding a replica of one ora plurality of chips of a back portion and a front portion of saidfinite length periodic sequence to the outside of said front portion andoutside of said back portion of said finite length period sequence,respectively, and demodulating with a receiver said extendedtransmission frame using a matched filter matched to said finite lengthperiod sequence prior to extension.
 2. The process fortwo-phase/four-phase modulated spread spectrum communication accordingto claim 1, further compromising: assigning different carrier waves torespective users, and assigning two-phase signals or four-phase signalsso that the comb-form modulated spectrum are generated on saidrespective carrier waves.
 3. The process for [A] two-phase/four-phasemodulated spread spectrum communication according to claim 1 or 2,further comprising forming said basic sequence as a two-phase orfour-phase orthogonal sequence or as a multi-phase orthogonal sequence,thereby anti-multi-path characteristics are reinforced by preventingside lobes from being generated in the vicinity of main pulses producedfrom said matched filter at said receiver.
 4. The process for [A]two-phase/four-phase modulated spread spectrum communication accordingto claim 1, further comprising preparing a plurality of different typesof said expanded transmission frame so as to be used as code words.